1. Field of the Invention
This invention relates generally to a linearly polarized microstrip antenna and more particularly to a microstrip antenna capable of obtaining a larger bandwidth and providing the desired frequency characteristics of impedance by loading a conductor plate.
2. Description of the Prior Art
Generally, a microstrip antenna utilizing an unbalanced plane circuit resonator provides various advantages such as being small in size, light in weight and low in height, but, on the other hand, it involves such disadvantages as having impedance characteristics which are, for example, narrower in bandwidth.
FIGS. 1 and 2 show an example of a circular microstrip antenna in the prior art. FIG. 1 is a plan view, while FIG. 2 is a cross-sectional view thereof. In these figures, reference numeral 1 denotes a patch or a radiating conductor element of an open-circuited plane circuit with the radius "a", number 2 represents a dielectric substrate (relative dielectric constant .epsilon..sub.r, thickness h) which is sufficiently thinner than the wavelength, 3 is a ground plane conductor and 4 is a feed point for feeding the radiating conductor element 1 from a coaxial line 5 at the input terminal through a feeder 6.
The operation is next explained. When energy is fed from the feed point 4, a wave is radiated from the open-circuit circumferential edge 7. The antenna shown in FIGS. 1 and 2 functions as a linearly polarized antenna.
The resonant angular frequency .omega..sub.0 in the fundamental mode of the resonator can be approximately as follows: EQU .omega..sub.0 .apprxeq..alpha..sub.c /a EQU .alpha.=1.841/.sqroot..epsilon..sub.r EQU c=3.times.10.sup.8 m/sec (the speed of light in a vacuum)
The bandwidth BW of the microstrip antenna is generally given as, EQU BW=(S-1)/(Q.sqroot.S)
(S is the standing wave ratio) and the bandwidth BW depends on the Q factor (quality factor).
Here, the Q factor of a microstrip antenna is given by the following relation: EQU 1/Q=1/Q.sub.r +1/Q.sub.d +1/Q.sub.c
Where, Q.sub.r is the Q factor corresponding to the radiation loss, Q.sub.d is the Q factor corresponding to the dielectric loss and Q.sub.c is the Q factor corresponding to the conductor loss.
These Q factors are approximately given by the following forms: EQU 1/Q.sub.d =tan .delta. (tan .delta. is the dielectric loss tangent of the dielectric substrate 2) EQU 1/Q.sub.c =.delta..sub.s /h(.delta..sub.s is the skin depth of the radiation conductor element 1 and the ground plane conductor 3) EQU 1/Q.sub.r =(4 .mu..sub.o ch/2.39 .lambda..sub.o).multidot.G.sub.r ##EQU1## Where .mu..sub.o =4.pi..times.10.sup.-7 H/m (the permeability in a vacuum) and .lambda..sub.o is the free space wavelength. G.sub.r is the equivalent conductance representing the radiation loss and when .epsilon..sub.r increases, G.sub.r reduces. Therefore, in the microstrip antenna, the increase of .epsilon..sub.r of the dielectric substrate 2 results in a rise in Q value, while the increase in thickness h of the substrate causes a reduction in Q value.
When the relative dielectric constant .epsilon..sub.r of the dielectric substrate 2 increases, energy is trapped within the circular microstrip antenna, causing the value of Q to become high and this results in a decrease in radiation efficiency. Thus, it is necessary, in order to improve the radiation efficiency and widen the bandwidth, to use a thicker substrate having a lower dielectric constant.
Here, it is noted that bidirectional communication is generally employed for public communications with the bandwidth of 8%-15%.
However, the microstrip antennas of the prior art use a low dielectric constant (for example, .epsilon..sub.r =2.2) substrate of teflon or polytetrafluoroethylene (PTFE) as the dielectric substrate 2 but it has a Q value less than 100. Such microstrip antennas have not been able to provide characteristics wherein the standing wave ratio is less than 1.5 over the required bandwidth of 8% or more even when the thickness of substrate is increased. Moreover, even in a case where the relative dielectric constant .epsilon..sub.r is lowered to about 1.1-1.3 by using, for example, a honeycomb substrate and the thickness of the substrate is increased with the intention of realizing a wider bandwidth, a standing wave ratio of 1.5 or less over the required bandwidth of 8% or more has not been obtainable.
Since microstrip antennas of the prior art are formed as described above, a problem has arisen in that the reflection characteristics are deteriorated in terms of the frequency characteristics of impedance when an antenna operates as a linearly polarized antenna with the bandwidth of 8% or more for public communication, etc.
Referring now to FIGS. 3 and 4, there is shown a rectangular microstrip antenna in the prior art. FIG. 3 is a plan view and FIG. 4 is a cross-sectional view wherein reference numeral 1 denotes a radiating conductor element of an open-circuited plane circuit of width "a" and length "b", number 2 denotes a dielectric substrate (relative dielectric constant .epsilon..sub.r, thickness h), 3 designates a ground plane conductor, and 4 designates a feed point for connecting a center conductor 6 of a coaxial cable 5 to the radiating conductor element 1.
The operation of the rectangular microstrip antenna is similar to the circular microstrip antenna shown in FIGS. 1 and 2 and when it is fed, it radiates a wave from an open-circuit circumferential edge 7 as a linearly polarized antenna.
The resonant frequency in the fundamental mode of the rectangular microstrip antenna is determined in a manner similar to the circular antenna. The resonant frequency is primarily determined by the length "b" of the radiating conductor element 1 and the relative dielectric constant .epsilon..sub.r of the dielectric substrate 2. On the other hand, the bandwidth primarily depends on the relative dielectric constant .epsilon..sub.r and the thickness h of the substrate 2. Although the bandwidth can be extended by making .epsilon..sub.r smaller and h thicker, the preferred range of the thickness h is limited in order to prevent the generation of a higher mode. The bandwidth of the microstrip antenna which has been put into practical use is not more than several percent.
The feed point impedance increases in value when the feed point 4 corresponds to the open-circuit circumferential edge 7, that is when m=0. Since the feed point impedance tends to decrease as the position of the feed point 4 shifts toward the center of the radiating conductor element 1, the dimension of "m" should be selected so as to match the feed point with the coaxial line 5 in impedance. At the same time, in order to prevent generation of cross-polarization, the dimension "n" should be a/2 (n=a/2).
Since the conventional microstrip antenna is constructed as above, the position of the coaxial line with respect to the radiating conductor element is limited to a particular point in view of the impedance matching and the suppression of the cross-polarization. Such problem becomes serious in terms of mounting antennas, in particular when a plurality of microstrip antennas are to be arranged at predetermined intervals to form an array antenna.
In addition, the conventional microstrip antenna has an essentially narrower bandwidth and deteriorates in reflection characteristics when applied for uses where a bandwidth of more than 8% is necessary, for example in public communication applications.